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LT1306 Synchronous, Fixed Frequency Step-Up DC/DC Converter
FEATURES
s s
DESCRIPTIO
s s s s s s s s
Output Disconnected from Input During Shutdown Output Voltage Remains Regulated When VIN > VOUT Controlled Input Current During Start-Up 300kHz Current Mode PWM Operation Can Be Externally Synchronized Internal 2A Switches Operates with VIN as Low as 1.8V Automatic Burst Mode Operation at Light Loads Quiescent Current: 160A Shutdown Current: 9A Typ
The LT(R)1306 is a fully integrated, fixed frequency synchronous boost converter capable of generating 5V at 1A from a Li-Ion cell. The device contains both the main power switch and synchronous rectifier on chip and automatically disconnects the output from the input in shutdown, eliminating the need for external load disconnect circuitry. Additionally, the output remains regulated when VIN exceeds VOUT, allowing difficult step-up/stepdown converter functions to be easily realized using a single inductor. The internal 300kHz oscillator of the LT1306 can be easily synchronized to an external clock from 425kHz to 500kHz. This allows switching harmonics to be tightly controlled and eliminates any beat frequencies that may result from a multifrequency system. The LT1306 automatically shifts into power saving Burst ModeTM operation at light loads. At heavy loads the LT1306 operates in fixed frequency current mode. No-load quiescent current is 160A and reduces to 9A in shutdown mode. The LT1306 is available in an SO-8 package.
APPLICATIO S
s s s s
Satellite Phones Portable Instruments Personal Digital Assistants Palmtop Computers
, LTC and LT are registered trademarks of Linear Technology Corporation. Burst Mode is a trademark of Linear Technology Corporation.
TYPICAL APPLICATION
D1 L1 10H C1 1F 90 85
EFFICIENCY (%)
1-CELL Li-Ion VIN S/S
SW LT1306
CAP OUT R1 768k FB GND R2 249k 5V 1A
+
CIN1 22F
CIN2 0.1F VC R3 118k CZ 68nF
+
CO1 220F
CO2 1F CIN1: AVX TAJC226M010 CO1: AVX TPSE227M010R0100 CIN1, CO2: CERAMIC C1: AVX TAJA105K020 D1: MMBD914LT1 L1: CTX10-2
CP 68pF
1306 F01
Figure 1. Single Li-Ion Cell to 5V Converter
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Efficiency
VIN = 4.2V VIN = 3.6V VIN = 2.6V 80 75 70 65 60 1 10 100 LOAD CURRENT (mA) 1000
1306 TA01
U
U
+
VO = 5V L1 = 10H (FIGURE 1)
1
LT1306
ABSOLUTE MAXIMUM RATINGS
(Note 1)
PACKAGE/ORDER INFORMATION
TOP VIEW VC 1 FB 2 VOUT 3 GND 4 8 S/S 7 VIN 6 CAP 5 SW
VIN Voltage ............................................................. 10V S/S Voltage ............................................................... 7V FB Voltage .............................................................. 10V VOUT Voltage .......................................................... 5.5V Junction Temperature .......................................... 125C Operating Temperature Range (Note 2) .. - 40C to 85C Storage Temperature Range ................. - 65C to 150C Lead Temperature (Soldering, 10 sec).................. 300C
ORDER PART NUMBER LT1306ES8
S8 PART MARKING 1306
S8 PACKAGE 8-LEAD PLASTIC SO
TJMAX = 125C, JA = 90C/ W
Consult factory for Industrial and Military grade parts.
The q denotes the specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25C. VIN = 2.5V, VS/S = VIN, VC open unless otherwise noted.
PARAMETER Reference Voltage Reference Line Regulation FB Input Bias Current Error Amplifier Transconductance Error Amplifier Output Source Current Error Amplifier Output Sink Current Error Amplifier Output Clamp Voltage VIN Undervoltage Lockout Threshold Idle Mode Output Leakage Current Output Source Current in Shutdown Switching Frequency Maximum Duty Cycle Switch Current Limit Burst Mode Operation Switch Current Limit Switch VCESAT Rectifier VCESAT Stepdown Mode Rectifier Voltage Switch and Rectifier Leakage Current ISW = 2A ISW = 2A VOUT = 0V, ISW = 1A VOUT = 2.2V, ISW = 1A VOUT = 0V, VIN = VSW = 7V, VCAP = 7.2V, VS/S = 0V
q
ELECTRICAL CHARACTERISTICS
CONDITIONS Measured at the FB Pin 1.8V VIN 7V VFB = VREF I = 0.2A VFB = 1V, VC = 0.8V VFB = 1.5V, VC = 0.8V VFB = 1V VFB = 1.5V, VOUT = 5.5V, VSW = 1.7V VOUT = 0V, VIN = VSW = 7V, VCAP = 7.2V, VS/S = 0V 1.8V VIN 7V, 0C TA 85C 1.8V VIN 7V, TA = - 40C VFB = 1V, 0C TA 85C VFB = 1V, TA = - 40C Duty Cycle = 0.1 (Note 3) Duty Cycle = 0.8 (Note 3)
q q q q q
MIN 1.22
TYP 1.24 0.002 10
MAX 1.26 0.1 25 220 11 11 1.38 1.8 15 -3 415 390
UNITS V %/V nA -1 A A V V A A kHz kHz % % A A
80 5 5 1.18 1.55
150 7.5 7.5 1.28 6
260 225 80 65 2.3 2.0
310 305 90 80
250 0.45 0.49 0.3 + VIN 1.3 0.1 0.575 0.675 0.7 + VIN 1.8 20
2
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W
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WW
W
mA V V V V A
LT1306
The q denotes the specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25C. VIN = 2.5V, VS/S = VIN, VC open unless otherwise noted.
PARAMETER S/S Pin Current Shutdown Pin Input High Voltage Shutdown Pin Input Low Voltage Shutdown Delay Synchronization Frequency Range Operating Supply Current Quiescent Supply Current Shutdown Supply Current CAP Pin Leakage Current Output Boost-to-Stepdown Threshold Output Stepdown-to-Boost Threshold Note 1: Absolute Maximum Ratings are those values beyond which the life to the device may be impaired. Note 2: The LT1306E is guaranteed to meet performance specifications from 0C to 70C. Specifications over the - 40C to 85C operating VS/S = VIN, VFB = 1.5V VS/S = 0V VIN = VCAP = 7V, VS/S = 2.5V, ISW = 0
q q
ELECTRICAL CHARACTERISTICS
CONDITIONS VS/S = VIN VS/S = 0V
MIN
TYP
MAX 6 -3
UNITS A A V V s kHz mA A A A V V
1.2 0.45 12 425 4.5 160 9 VIN VIN - 0.1 20 50 500 8 250 16 10
temperature range are assured by design, characterization and correlation with statistical process controls. Note 3: Switch current limit guaranteed by design/correlation to static tests.
TYPICAL PERFORMANCE CHARACTERISTICS
Maximum Load Current vs Input Voltage
1.5 VO = 5V
REFERENCE VOLTAGE (V)
VO = 3.3V
ILOADMAX (A)
1.0
IS/S (A)
0.5
L = 10H TJ = 125C
0 1.5 2.0 2.5 3.0 3.5 4.0 4.5 5.0 5.5 6.0 6.5 7.0 VIN (V)
1306 * G01
UW
TA = 25C TA = 50C
Reference Voltage vs Temperature
1.239 1.238 1.237 1.236 1.235 1.234 1.233 1.232 1.231 -40 -20 0 20 40 60 80 100 5 4 3 2 1 0 -1 -2 -3 -4 -5
S/S Pin Current vs S/S Pin Voltage
TA = -40C TA = 25C TA = 85C
0
1
TEMPERATURE (C)
1306 * G02
2 3 VS/S (V)
4
5
1306 * G03
3
LT1306 TYPICAL PERFORMANCE CHARACTERISTICS
Shutdown Supply Current vs Input Voltage
40 35
SUPPLY CURRENT (A)
5.0
TA = 25C
S/S CURRENT (A)
VS/S = 2.5V 2.5
IDLE-MODE SUPPLY CURRENT (A)
30 25 20 15 TA = -40C 10 5 0 2 4 6 8 INPUT VOLTAGE (V) 10 12
1306 * G04
TA = 85C
Oscillator Frequency Line Regulation
320
315 310 305
315
FREQUENCY (kHz) FREQUENCY (kHz)
295 290 285 280 275
DUTY RATIO (%)
310
305
300 0 1 2 3 4 5 VIN (V)
1306 * G07
6
7
Maximum Allowable Rise Time of Synchronizing Pulse
600 500 3.0
MAXIMUM RISE TIME (ns)
SWITCH VOLTAGE (V)
400 300 200 100 0
CURRENT LIMIT (A)
1
3.5 1.5 2.0 2.5 3.0 SYNCHRONIZING PULSE AMPLITUDE (V)
1306 * G10
4
UW
8 9 10
S/S Pin Current vs Temperature
155
Idle-Mode Supply Current vs Temperature
150
145
0 VS/S = 0V -2.5 - 40
140
-20
0 20 40 60 TEMPERATURE (C)
80
100
135 -40
-20
0
20
40
60
80
100
TEMPERATURE (C)
1306 * G05
1306 * G06
Frequency vs Temperature
95 90 85 80 75 70 65 270 265 -40 -20 40 0 60 20 TEMPERATURE (C) 80 100
Maximum Duty Ratio
VIN = 2.5V
300
60 -40 -20
40 20 60 0 TEMPERATURE (C)
80
100
1306 * G08
1306 * G09
Current Limit vs Duty Cycle
0.7 TA = 25C 0.6 2.8 0.5 0.4
Switch Saturation Voltage vs Current
TA = 25C
2.6
TA = 85C 0.3 0.2 0.1 0
TA = -40C
2.4
2.2
2.0 0 10 20 30 40 50 60 DUTY CYCLE (%) 70 80 90
0
0.5
1.0 2.0 1.5 SWITCH CURRENT (A)
2.5
1306 * G12
1306 * G11
LT1306 TYPICAL PERFORMANCE CHARACTERISTICS
Rectifier Saturation Voltage vs Current
0.7 0.6
RECTIFIER VOLTAGE (V)
1.90
TA = 85C
RECTIFIER VOLTAGE (V)
0.5 0.4 TA = -40C 0.3 0.2 0.1 0 0 0.5 2.0 1.5 RECTIFIER CURRENT (A) 1.0 2.5
1306 * G13
TA = 25C
Start-Up to Shutdown Transient Response*
VS/S 5V/DIV VSW 5V/DIV IL 2A/DIV VSW 5V/DIV
VO 5V/DIV
VIN = 2.5V
1ms/DIV
*Notice that the Input Start-Up Current is well Controlled and the Output Voltage Falls to Zero in Shutdown.
UW
Stepdown-Mode Rectifier Voltage vs Current
1.85 1.80 1.75 1.70 1.65 1.60 1.55 0 0.5 1.0 1.5 RECTIFIER CURRENT (A) 2.0
1306 * G14
Continuous-Conduction Mode Switching Waveforms in Boost Operation
VIN = 6V VOUT = 5V TA = 25C
VSW 5V/DIV IL 0.5A/DIV
VO 0.1V/DIV AC VIN = 4.2V VO = 5V 2s/DIV
Continuous-Conduction Mode Switching Waveforms in Stepdown Mode
LOAD CURRENT 0.5A/DIV DC INDUCTOR CURRENT 1A/DIV
Transient Response of the Converter in Figure 1 with a 50mA to 800mA Load Step
IL 0.5V/DIV VO 50mV/DIV AC
OUTPUT 0.1V/DIV AC VIN = 6V VO = 5V 2s/DIV VIN = 3.6V VO = 5V 1ms/DIV
5
LT1306
PIN FUNCTIONS
VC (Pin 1): Compensation Pin for Error Amplifier. VC is the output of the transconductance error amplifier. Loop frequency compensation is done by connecting an RC network from the VC pin to ground. FB (Pin 2): Inverting Input of the Error Amplifier. Connect the resistor divider tap here. Set output voltage according to VOUT = 1.24V (1 + R1/R2). VOUT (Pin 3): Output of the Switching Regulator and Emitter of the Synchronous Rectifier. Connect appropriate output capacitor from here to ground. VOUT must be kept below 5.5V. GND (Pin 4): Ground. Connect to local ground plane. SW (Pin 5): Switch Pin. The collectors of the grounded power switch and the synchronous rectifier. Keep the SW trace as short as possible to minimize EMI. CAP (Pin 6): Power Supply to the Synchronous Rectifier Driver. The bootstrap capacitor and the blocking diode are tied to this pin. The CAP voltage switches between a low level of VIN - VD to a high level determined by the VSW high level. VIN (Pin 7): Supply or Battery Input Pin. Must be closely bypassed to ground plane. S/S (Pin 8): Shutdown and Synchronization Pin. Shutdown is active low with a typical threshold of 0.9V. For normal operation, the S/S pin is tied to VIN. To externally synchronize the switching regulator, drive the S/S pin with a pulse train.
BLOCK DIAGRA
1.24V
+
A1 gm
FB 2
-
VB
300kHz OSC S/S 8 SYNC
CLK PWM CONTROL
SHDN
REF/BIAS
SHUTDOWN DELAY
Figure 2. LT1306 Block Diagram
6
-
A2 SENSE AMP
+
RAMP COMPENSATION
W
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VC 1
VIN 7
-
A5 UVLO IRECT > 0 CAP 6 X3 A3 X4 IDLE X5
1.65V
+ - +
DCM CONTROL
5 SW
-
A4 S
X1 Q2 Q R X2 X4 Q1 RECTIFIER IRECT + VCE2 -
OUT 3
+ +
+
RS
4 GND
1306 F02
LT1306
OPERATIO
The LT1306 is a fixed frequency current mode PWM regulator with integrated power transistor Q1 and synchronous rectifier Q2. In the Block Diagram, Figure 2, the PWM control circuit is enclosed within the dashed line. It consists of the current sense amplifier (A2), the oscillator, the compensating ramp generator, the PWM comparator (A4), the logic (X1 and X2), the power transistor driver (X4) and the main power switch (Q1). Notice that the clock (CLK) "blanks" Q1 conduction. The internal oscillator frequency is 300kHz. The pulse width of the clock determines the maximum on duty ratio of Q1. In the LT1306 this is set to 88%. Q1 turns on at the trailing edge of the clock pulse. To prevent subharmonic oscillation above 50% duty ratio, a compensating ramp (generated from the oscillator sawtooth) is added to the sensed Q1 current. Q1 is turned off when this sum exceeds the error amplifier A1 output, VC. Q1's absolute current limit is reached when VC's upward excursion is clamped internally at 1.28V. The error amplifier output, VC, determines the peak switch current required to regulate the output voltage. VC is a measure of the output power. At heavy loads, the average and the peak inductor currents are both high. VC moves to the upper end of its operating range and the LT1306 operates in continuous conduction mode (CCM). As load decreases, the average inductor current decreases. In CCM, the peak-to-peak inductor current ripple to the first order depends only on the inductance, the input and the output voltages. When the average inductor current falls below 1/2 of the peak-to-peak inductor current ripple, the converter enters discontinuous conduction mode (DCM). The switching frequency remains constant except that the inductor current always returns to zero within each switching cycle. In both CCM and DCM, the output voltage is regulated with negative feedback. A1 amplifies the error voltage between the internally generated 1.24V reference and the attenuated output voltage. The RC network from the VC pin to ground provides the loop compensation. Further reduction in the load moves VC towards the lower end of its operating range. Both the peak inductor current
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and switch Q1's on-time decrease. Hysteretic comparator A3 determines if VC is too low for the LT1306 to operate efficiently. As VC falls below the trip voltage VB, the output of A3 goes high. All circuits except the error amplifier, comparators A3 and A5, and the rectifier driver control X5, are turned off. After the remaining energy stored in the inductor is delivered to the output through the synchronous rectifier Q2, the LT1306 stops switching. In this idle state, the LT1306 draws only 160A from the input. With switching stopped and the load being powered by the output filter capacitor, the output voltage decreases. VC then starts to increase. Q1 does not start to switch until VC rises above the upper trip point of A3. The LT1306 again delivers power to the output as a current mode PWM converter except that the switch current limit is only about 250mA due to the low value of VC. If the load is still light, the output voltage will rise and VC will fall, causing the converter to idle again. Power delivery therefore occurs in bursts. The on-off cycle frequency, or burst frequency, depends on the operating conditions, the inductance and the output filter capacitance. The output voltage ripple in Burst Mode operation is usually higher than either CCM or DCM operation. Burst Mode operation increases light load efficiency because it delivers more energy to the output during each clock cycle than is possible with DCM operation's extremely low peak switch current. This allows fewer switching cycles per unit time to maintain a given output. Chip supply current therefore becomes a small fraction of the total input current. The synchronous rectifier is represented as NPN transistor, Q2, in the Block Diagram (Figure 2). A rectifier drive circuit, X5, supplies variable base drive to Q2 and controls the voltage across the rectifier. The supply voltage, VCAP, for the driver is generated locally with the bootstrap circuit, D1 and C1 (Figure 1). When Q1 is on, the bootstrap capacitor C1 is charged from the input to the voltage VIN - VD1(ON) - VCESAT1. The charging current flows from the input through D1, C1 and Q1 to ground. After Q1 is switched off, the node SW goes above VO by the rectifier drop VCESAT2. D1 becomes back-biased and the CAP voltage is pushed up to VO + VCESAT2 + VIN - VD1(ON) - VCESAT1. C1 supplies the base drive to Q2. The consumed charge is replenished during the Q1 on interval.
7
LT1306
OPERATIO
In boost operation, X5 drives the rectifier Q2 into saturation. The voltage across the rectifier is VCESAT. As the inductor current decreases, Q2's base drive also decreases. X5 ceases supplying base current to Q2 when the inductor current falls to zero. If VIN > VO, Q2 will no longer be driven into saturation. Instead the voltage across Q2 is allowed to increase so that the inductor voltage reverses polarity as Q1 switches. Since the inductor voltage is bipolar, volt-second balance can be maintained regardless of the input voltage. The LT1306 is therefore capable of operating as a step-down regulator with the basic boost topology. Input start-up current is also well controlled since the inductor current cannot increase during Q1's off-time with negative inductor voltage. The rectifier voltage drop depends on both the input and the output voltages. Efficiency in the step-down mode is less than that of a linear regulator. For sustained stepdown operation, the maximum output current will be limited by the package thermal characteristics.
MODE BOOST
STEPDOWN
1306 F03
0
Figure 3. DC Transfer Characteristics of the Mode Control Comparator Plotted with VO as an Independent Variable. VIN is Considered Fixed.
8
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A hysteretic comparator in driver X5 controls the mode of operation. DC transfer characteristics of the comparator are shown in Figure 3 and Figure 4. A logic low at the S/S pin (Pin 8) initiates shutdown. First, all circuit blocks in the LT1306 are switched off. The synchronous rectifier Q2 and its driver are kept on to allow stored inductive energy to flow to the output. As VO drops below VIN, the voltage across the rectifier Q2 increases so that the inductor voltage reverses. Inductor current continues to fall to zero. Driver X5 then turns off and the rectifier, Q2, becomes an open circuit. The LT1306 dissipates only 9A in shutdown. The LT1306 is guaranteed to start with a minimum VIN of 1.8V. Comparator A5 senses the input voltage and generates an undervoltage lockout (UVLO) signal if VIN falls below this minimum. In UVLO, VC is pulled low and Q1 stops switching. The LT1306 draws 160A from the input.
MODE BOOST STEPDOWN
1306 F04
VIN - 0.1V VIN
VO
VO
VO + 0.1V
VIN
Figure 4. DC Transfer Characteristics of the Mode Control Comparator Plotted with VIN as an Independent Variable. VO is Considered Fixed.
LT1306
APPLICATIONS INFORMATION
Output Voltage Setting The output voltage of the LT1306 is set with a resistive divider, R1 and R2 (Figure 1 and Figure 5), from the output to ground. The divider tap is tied to the FB pin. Current through R2 should be significantly higher than the FB pin input bias current ( 25nA). With R2 = 249k, the input bias current of the error amplifier is 0.5% of the current in R1.
VO R1 FB PIN R2 VO = 1.24V 1 + R1 R2 VO -1 R1 = R2 1.24
() ()
1306 F05
Figure 5. Feedback Resistive Divider
Synchronization and Shutdown The S/S pin (Pin 8) can be used to synchronize the oscillator or disconnect the load from the input. The S/S pin is tied to the input (VIN > 1.8V) for normal operation. The oscillator in the LT1306 can be externally synchronized by driving the S/S pin with a pulse train (See the graph "Maximum Allowable Rise Time of Synchronizing Pulse" in the Typical Performance Characteristics). The synchronization is positive edge triggered. The recommended frequency of the external clock ranges from 425kHz to 500kHz. If synchronization results in switching jitter, reducing the rising edge dv/dt of the external clock pulse usually cures the problem. Shutdown will be activated if the S/S pin voltage stays below the shutdown threshold (0.45V) for more than 50s. This shutdown delay is reset whenever the S/S pin goes above the shutdown threshold. Inductor The value of the energy storage inductor L1 (Figure 1) is usually selected so that the peak-to-peak ripple current is less than 40% of the average inductor current. For 1- or 2-cell alkaline or single Li-Ion to 5V applications, 10H to 20H is recommended for the LT1306 running at 300kHz. A 5H to 10H inductor can be used if the LT1306 is externally synchronized at 500kHz.
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The inductor should be able to handle the full load peak inductor current without saturation. The peak inductor current can be as high as 2A. This places a lower limit on the core size of the inductor. Powder iron cores have unacceptable core losses and are not suitable for high efficiency applications. Most ferrite core materials have manageable core losses and are recommended. Inductor DC winding resistance (DCR) also needs to be considered for efficiency. Usually there are trade-offs between core loss, DCR, saturation current, cost and size. For EMI sensitive applications, one may want to use magnetically shielded or toroidal inductors to contain field radiation. Table 1 lists a number of inductors suitable for LT1306 applications.
Table 1. Inductors Suitable for Use with the LT1306
VENDOR Coilcraft PART NO. DO3308-103 DO3316-472 DO3316-103 DO3316-153 Coiltronics Murata Sumida CTX5-2 CTX10-2 LQN6C4R7 CDRH73-100 CD43-4R7 VALUE MAX DCR (H) () 5.0 10 4.7 10 15 5 10 4.7 10 4.7 0.023 0.09 0.018 0.029 0.046 0.021 0.032 0.034 0.072 0.109 CORE TYPE Toroid Open Open Open Open Toroid Toroid Open Magnetic Shielding Open HEIGHT (mm) 4.8 3.0 5.2 5.2 5.2 6.0 6.0 5.0 3.4 3.2
BH Electronics 511-0033
Capacitors The output filter capacitor is usually chosen based on its equivalent series resistance (ESR) and the acceptable change in output voltage as a result of load transients. The output voltage ripple at the switching frequency can be estimated by considering the peak inductor current and the capacitor ESR.
IPEAK IIN
(IO)(VO )
VIN
output ripple (ESR)(IPEAK) =
(ESR)(IO )(VO )
VIN
9
LT1306
APPLICATIONS INFORMATION
Since a boost converter produces high output current ripple, one also needs to consider the maximum ripple current rating of the output capacitor. Capacitor reliability will be affected if the ripple current exceeds the maximum allowable ratings. This maximum rating is usually specified as the RMS ripple current. In the LT1306 the RMS output capacitor ripple current is: switch and can cause the current limit comparator to trip erratically. For boost applications where VIN is a few tenths of a volt below VO, a 1F or 2.2F tantalum capacitor (such as AVX TAJ series) can be used for C1. The ESR of the tantalum capacitor limits the charging current. A low value resistor (2 to 5) can also be added in series with C1 for further limiting the charging current although this tends to lower the converter efficiency slightly. Frequency Compensation Current mode switching regulators have two feedback loops. The inner current feedback loop controls the inductor current in response to the outer loop. The outer or overall feedback loop tightly regulates the output voltage. The high frequency gain asymptote of the inner current loop rolls off at - 20dB/decade and crosses the unity gain axis at a frequency c between 1/6 to 2/3 of the switching frequency. The current loop is stable and is wideband compared to the overall voltage feedback loop. The low frequency current loop gain is not high (usually between unity and 10) but it increases the low frequency impedance of the inductor as seen by the output filter capacitor. (In a boost regulator, the inductor is connected to the output during the switch off-time.) Current mode control introduces an effective series resistance (>> DCR) to the inductor that damps the LC tank response. The complex high-Q poles of the LC filter are now separated, resulting in a dominant pole determined by the filter capacitance and the load resistance and a second high frequency pole. For a boost regulator the control to output transfer function can be shown to have a dominant pole at the load corner frequency P = 1 RL (CO ) 2
IO
VO - VIN VIN
For 2-cell to 5V applications, 220F low ESR solid tantalum capacitors (AVX TPS series or Sprague 593D series) work well. To reduce output voltage ripple due to heavy load transients or Burst Mode operation, higher capacitance may be used. For through-hole applications, Sanyo OS-CON capacitors are also good choices. In a boost regulator, the input capacitor ripple current is much lower. Maximum ripple current rating and input voltage ripples are not usually of concern. A 22F tantalum capacitor soldered near the input pin is generally an adequate bypass. Bootstrap Supply Diode D1 and capacitor C1 generate a pulsating supply voltage, VCAP, which is higher than the output. The rectifier drive circuit runs off this supply. During rectifier on-time, the rectifier base current drains C1. Q2 base current and the maximum allowable VCAP ripple voltage determine the size of C1. A 1F capacitor is sufficient to keep VCAP ripple below 0.3V. For a 2-cell input (VIN > 1.8V) over an extended temperature range, a BAT54 Schottky diode may be used for D1. The use of a Schottky diode increases the bootstrap voltage and the operating headroom for the rectifier driver, X5. Diodes like a 1N4148 or 1N914 work well for 2-cell inputs over the 0C to 70C commercial temperature range. The charge drawn from C1 during the rectifier on-time has to be replenished during the switch on-interval. As duty cycle decreases, the amplitude of the C1 charging current can increase dramatically especially when delivering high power to the load. This charging current flows through the
10
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and a moving right-half plane (RHP) zero with a minimum value of R (1 - DMAX ) Z = L L
2
LT1306
APPLICATIONS INFORMATION
where RL = Maximum Load = DMAX Output Voltage Maximum DC Load Current = Maximum Converter Duty Cycle The low frequency zero 1/R3CZ of the compensation network is placed at P/2. CZ = 2 R3P
=
VO - VIN(MIN) + 0.5 VO + 0.1
There is also a second pole at the current loop crossover frequency C (Figure 6). Z is much lower in frequency than C. The loop is compensated by adjusting the midband gain with resistor R3 (Figure 7) so that the overall loop gain crosses 0dB before the minimum frequency RHP zero (i.e., corresponding to the highest duty ratio). The value of R3 can be estimated with the fromula: R3 = 390 VO(1 - DMAX )CORL L
Due to the low transconductance of the error amplifier, the gain setting resistor R3 is AC-coupled with capacitor CZ. This prevents R3 from inducing an offset to the input of the error amplifier. It also creates a pole at DC and a low frequency zero. The amplitude response of the error amplifier with the compensation network shown is:
1 + S * R3 * CZ R2 VC = gm VO R1 + R2 S * CZ 1 + S * R3 * CP CZ >> CP
[(
(
)
)]
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The capacitor CP ensures adequate gain margin beyond the RHP zero. The high frequency pole 1/R3CP of the amplifier frequency response is placed beyond Z. CP = 1 3 ZR3
Higher output filter capacitance rolls off the gain response from a lower corner frequency so higher midband gain is required in the compensation network to make the overall loop gain cross 0dB just below Z. Layout Consideration To minimize EMI and high frequency resonances, it is essential to keep the SW and the CAP trace leads as short as possible. The input and the output bypass capacitors CIN and COUT should be placed close to the IC package and soldered to the ground plane. A ground plane under the switching regulator is highly recommended. Figure 8 shows a suggested component placement and PC board layout.
11
LT1306
APPLICATIONS INFORMATION
GAIN (dB) MIDBAND GAIN = (gm)(R3)(R2) R1 + R2
AMOUNT OF MIDBAND GAIN NEEDED P 1 1 (R3)(CZ) RL (CO) 2 Z C
0
()
LOOP GAIN CROSSOVER FREQUENCY 1 Z 3
AMPLITUDE RESPONSE OF CONTROL-TO-OUTPUT TRANSFER FUNCTION BEFORE COMPENSATION
VO VC Figure 6. Gain Asymptotes of the Control-to-Output and Error Amplifier Transfer Function VC VO
L VIN Q2 PWM CONTROL LOGIC VO
gm
LT1306 VC R3 CZ CP
GND
Figure 7. Current Mode Boost Converter Overall-Loop Compensation
12
-
+
U
W
U
U
VC V
O
R (1 - DMAX)2 RHP ZERO = L L AMPLITUDE RESPONSE OF THE ERROR AMPLIFIER CURRENT LOOP CROSSOVER FREQUENCY
OVERALL LOOP GAIN AFTER COMPENSTION
VO V
C
1306 F06
SW
Q1 RECTIFIER FB
R1 IO
1.24V
R2
CO
RL
1306 F07
LT1306
APPLICATIONS INFORMATION
GROUND PLANE CZ
R2 R1
VOUT
CO1 VIAS
Figure 8. Recommended Component Placement for LT1306. Notice That the Input and the Output Capacitors Are Grounded at the Same Point. A Ground Plane Under the DC/DC Converter Is Highly Recommended. Use Multiple Vias to Tie Pin 4 Copper to the Ground Plane
U
+
W
U
U
R3 CP VC 1 2 3 4 CIN2 LT1306 8 7 6 5
S/S VIN
D1
C1
+
+
CO2
CIN1
GND
L1
1306 F08
13
LT1306
TYPICAL APPLICATIO S U
2-Cell NiMH to 3.3V Output
D1 L1 4.7H 2V TO 3V 2V/500kHz VIN S/S CIN1 0.1F CERAMIC SW LT1306 CIN2 22F VC R3 95k CZ 5.6nF FB GND R2 249k C1 1F
+
CP 39pF
Efficiency
90 85
EFFICIENCY (%)
VO = 3.3V L1 = 4.7H VIN = 3V VIN = 2.5V VIN = 1.8V
80 75 70 65 60 1
10 100 LOAD CURRENT (mA)
14
+
CAP OUT R1 412k 3.3V 1A
+
CO 220F CIN1: AVX TAJC226M010 CO1: AVX TPSE227M010R0100 C1: AVX TAJA105K020 D1: CMDSH-3 L1: LQN6C4R7
1306 F09
1000 2000
1306 F09a
LT1306
PACKAGE DESCRIPTION
0.010 - 0.020 x 45 (0.254 - 0.508) 0.008 - 0.010 (0.203 - 0.254) 0- 8 TYP
0.014 - 0.019 (0.355 - 0.483) TYP *DIMENSION DOES NOT INCLUDE MOLD FLASH. MOLD FLASH SHALL NOT EXCEED 0.006" (0.152mm) PER SIDE **DIMENSION DOES NOT INCLUDE INTERLEAD FLASH. INTERLEAD FLASH SHALL NOT EXCEED 0.010" (0.254mm) PER SIDE
0.016 - 0.050 (0.406 - 1.270)
Information furnished by Linear Technology Corporation is believed to be accurate and reliable. However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights.
U
Dimensions in inches (millimeters) unless otherwise noted.
S8 Package 8-Lead Plastic Small Outline (Narrow 0.150)
(LTC DWG # 05-08-1610)
0.189 - 0.197* (4.801 - 5.004) 8 7 6 5
0.228 - 0.244 (5.791 - 6.197)
0.150 - 0.157** (3.810 - 3.988)
1
2
3
4
0.053 - 0.069 (1.346 - 1.752)
0.004 - 0.010 (0.101 - 0.254)
0.050 (1.270) BSC
SO8 1298
15
LT1306
TYPICAL APPLICATIO S
4-Cell NiMH to 5V Output
D1 L1 10H C1 1F 90 85 VO = 5V L1 = 10H
VIN S/S
SW LT1306
CAP OUT R1 768k FB GND R2 249k 5V 1A
EFFICIENCY (%)
3.6V TO 6.5V
+
CIN1 22F
CIN2 0.1F CERAMIC R3 75k CZ 15nF
VC
CP 22pF
RELATED PARTS
PART NUMBER LT1302 LT1304 LT1307/LT1307B LT1308A/LT1308B LT1316 LT1317/LT1317B LT1610 LT1613 LT1615 LTC1624 LT1949 DESCRIPTION High Output Current Micropower DC/DC Converter 2-Cell Micropower DC/DC Converter Single Cell, Micropower, 600kHz PWM DC/DC Converters High Output Current Micropower DC/DC Converter Burst Mode Operation DC/DC with Programmable Current Limit Micropower, 600kHz PWM DC/DC Converters Single-Cell Micropower DC/DC Converter 1.4MHz Switching Regulator in 5-Lead SOT-23 Micropower Step-Up DC/DC in 5-Lead SOT-23 High Efficiency N-channel Switching Regulator Controller 600kHz, 1A Switch PWM DC/DC Converter COMMENTS 5V/600mA from 2V, 2A Internal Switch, 200A IQ 5V/200mA, Low-Battery Detector Active in Shutdown 3.3V at 75mA from One Cell, MSOP Package 5V at 1A from Single Li-Ion Cell 1.5V Minimum, Precise Control of Peak Current Limit 100A IQ, Operate with VIN as Low as 1.5V 3V at 30mA from 1V, 1.7MHz Fixed Frequency 5V at 200mA from 4.4V Input, Tiny SOT-23 package 20A IQ, 36V/350mA Internal Switch, VIN as Low as 1.2V VOUT = 1.19V to 30V in Stepdown; VIN = 3.5V to 36V SO-8 Package 1.1A, 0.5/30V Internal Switch, VIN as Low as 1.5V
1306f LT/TP 0400 4K * PRINTED IN USA
16
Linear Technology Corporation
1630 McCarthy Blvd., Milpitas, CA 95035-7417
(408)432-1900 q FAX: (408) 434-0507 q www.linear-tech.com
U
+
Efficiency
VIN = 4.8V
80 75 70 65 60 1
VIN = 3.6V
VIN = 6V
+
CO1 220F
CO2 1F CERAMIC CIN1: AVX TAJC226M010 CO1: AVX TPSE227M010R0100 C1: AVX TAJA105K020 D1: MMBD914LT1 L1: CTX10-3
10 100 LOAD CURRENT (mA)
1000 2000
1306 F10a
1306 F09
Transient Response with Step Input (4V to 6V)
VIN 5V/DIV VSW 5V/DIV
IL 500mA/DIV VO 0.1V/DIV AC 0.5ms/DIV
(c) LINEAR TECHNOLOGY CORPORATION 1999

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